To increase the amount of information that can be transmitted in a given bandwidth of the radio frequency (RF) spectrum, i.e., to increase what is referred to in the art as “spectral efficiency,” state-of-the-art and evolving wireless communications technologies, such as the Wideband Code Division Multiple Access (W-CDMA) air interface used in third generation (3G) Universal Mobile Telecommunications System (UMTS) cellular networks, and the Evolved UMTS Terrestrial Radio Access (E-UTRA) air interface proposed for use in the fourth generation (4G) Long Term Evolution (LTE) UMTS cellular network upgrade, employ nonconstant-envelope modulation schemes.
Although more spectrally efficient than constant-envelope modulation schemes, nonconstant-envelope modulation schemes require that the output power of conventional communications transmitters (i.e., those transmitters based on quadrature modulators) to be backed off in order to prevent signal distortion. Failure to back-off the power causes the peaks of the nonconstant-envelope signals to be undesirably clipped. The resulting distortion reduces the modulation accuracy of the transmitter and makes it difficult to comply with communications standards and governmental regulations.
Power back-off in a transmitter is achieved by biasing the transmitter's power amplifier (PA) so that the PA is forced to operate in its linear region of operation for the full range of output powers the PA must be configured to operate. In a typical transmitter design, this is achieved by biasing the PA so that its peak output power does not exceed the PA's 1-dB compression point, which defines the input power at which the gain of the PA drops by 1 dB from its ideal, linear response value. The degree of power back-off required depends on the peak-to-average ratio (PAR) of the nonconstant-envelope signals applied to the input of the PA. The PAR is determined by the modulation scheme being used. The higher the PAR, the more the output power of the PA must be backed off. For example, for a PAR of 3 dB, an average power of 10 dBm and a peak output power of 13 dBm, a linear PA response requires that the average output power of the PA be backed off by at least 3 dB in order for the peak output power not to exceed the 1-dB compression point.
Employing power back-off does help to ensure PA linearity. However, it also undesirably results in a significant reduction in energy efficiency. The energy efficiency of a transmitter is determined in large part by how efficient the transmitter's PA is, since the PA is usually the most energy-consuming component of the transmitter. The efficiency of a PA is defined as the ratio of the PA RF output power to the direct current (DC) power supplied to the PA. Efficiency is therefore high when the PA is operating at a high RF output power, but low when the PA is operating at low RF output powers. In most any practical application, the PA operates at high or peak RF output powers only for very short periods of time. For all other times, the RF output power is backed off and the efficiency of the PA is, as a consequence, substantially lowered.
The low energy efficiency of conventional quadrature-modulator-based transmitters is a major problem, particularly in mobile handset applications since the transmitter and its PA are powered by a battery. Fortunately, a more efficient type of communications transmitter, known as a polar transmitter, is available. In a polar transmitter, the amplitude information (i.e., the signal envelope) is temporarily removed from the nonconstant-envelope signal so that the polar transmitter's PA can be operated in its nonlinear region, where it is more efficient at converting energy from the transmitter's power supply into RF power than it is when configured to operate in its linear region. As explained in more detail below, the signal envelope is restored at the output of the PA by dynamically controlling the PA's power supply according to amplitude variations in the signal envelope.
FIG. 1 is a drawing showing the basic elements of a polar transmitter 100. The polar transmitter 100 includes a digital signal processor (DSP) 102; a Coordinate Rotation Digital Computer (CORDIC) converter 104; an amplitude modulation (AM) path including a first digital-to-analog converter (DAC) 106 and an amplitude modulator 108; a phase modulation (PM) path including a second DAC 110 and a phase modulator 112; a PA 114; and an antenna 116.
The DSP 102 operates to generate rectangular-coordinate in-phase and quadrature phase (i.e., I and Q) signals from bits in a digital message to be transmitted. The DSP 102 formats the I and Q signals in accordance with a predetermined modulation scheme, pulse-shapes the I and Q signals to reduce signal bandwidth, and then couples the pulse-shaped I and Q signals to inputs of the CORDIC converter 104. The CORDIC converter 104 converts the pulse-shaped I and Q signals into a digital amplitude component signal ρ representing the envelope of the modulation and a digital phase component signal Δθ representing the sample time by sample time phase difference of the modulation.
The first and second DACs 106 and 110 convert the digital amplitude and phase component signals ρ and Δθ into analog amplitude and phase difference modulation signals, which are coupled to inputs of the amplitude modulator 108 and the phase modulator 112, respectively. The amplitude modulator 108 operates to modulate a DC power supply Vsupply according to amplitude variations in the analog amplitude modulation signal, to generate an amplitude-modulated power supply signal Vs(t), which is coupled to the power supply port of the PA 114. Meanwhile, the phase modulator 112 operates to modulate an RF carrier signal according to the changes in phase in the analog phase difference modulation signal, to generate a phase-modulated RF carrier signal RFin, which is coupled to the RF input port of the PA 114.
Because the phase-modulated RF carrier signal RFin has a constant envelope, the PA 114 can be configured to operate in its nonlinear region of operation without the risk of signal peak clipping. Typically, the PA 114 is implemented as a Class D, E or F switch-mode PA 114 operating between compressed and cut-off states, so that the output power of the PA 114 is directly controlled and modulated according to the amplitude variations in the amplitude-modulated power supply signal Vs(t). By modulating the power supply port of the PA 114 in this manner, the amplitude modulation represented in the original digital amplitude component signal ρ is restored at the output of the PA 114, as the PA 114 amplifies the phase-modulated RF carrier signal RFin.
Although the polar transmitter 100 is capable of processing and transmitting nonconstant-envelope signals at high efficiencies, recomposition of the AM and PM information at the output of the PA 114 can be difficult due to the fact that the AM and PM are processed independently, i.e., in the separate AM and PM paths. The different levels of signal processing and difference in delays presented to signals in the AM and PM paths results in a misalignment (i.e., a delay mismatch) of the signals arriving at the power supply and RF input ports of the PA 114. Failure to reduce the delay mismatch degrades the modulation accuracy of the polar transmitter 100 and undesirably leads to the generation of out-of-band signal energy at the output of the PA 114, making it difficult, or in some cases even impossible, to comply with communications standards. For example, in transmitters configured to operate in according with the W-CDMA air interface in a Universal Mobile Telecommunications System (UMTS), compliance in adjacent channels 206a and 206b requires that the power spectral density (PSD) of the output of the transmitter's PA fall below a spectral mask 202, as illustrated in FIG. 2. The energy generated by the polar transmitter 100 in adjacent channels 206a and 206b increases as the delay mismatch increases. If the delay mismatch is too high, the spectral mask requirements cannot be satisfied.
The amount of delay mismatch between the AM and PM paths of the polar transmitter 100 that may be tolerated in any given application depends on the type of modulation being employed. For example, in Enhanced Data Rates for Global System for Mobile Communications (GSM) Evolution (EDGE) systems, a delay mismatch of up to about 10 ns may be tolerable. However, in wideband modulation systems, such as in W-CDMA systems, where higher clock rates are required to preserve the much wider modulation signal bandwidths, the delay mismatch should be reduced to 1 ns or less.
Delay mismatch in the polar transmitter 100 is also affected by integrated circuit process, voltage and temperature (PVT) variations. In narrowband applications such as GSM and EDGE, PVT variations on delay mismatch are small enough that they can be either ignored or compensated for by inserting a fixed compensating delay in either of the AM or PM paths during a preoperational mode calibration process performed during initial system setup. In wideband applications, however, such as W-CDMA and next-generation wideband applications, the PVT variations are strong enough that they can neither be ignored nor sufficiently compensated for in preoperational mode calibration processes.
It would be desirable, therefore, to have methods and apparatus for dynamically monitoring and adaptively reducing the delay mismatch between the AM and PM paths of a polar transmitter.